Entertaining experiments: some possibilities of a field-effect transistor

Radio magazine, number 11, 1998.

It is known that the input resistance of a bipolar transistor depends on the load resistance of the cascade, the resistance of the resistor in the emitter circuit and the base current transfer coefficient. Sometimes it can be relatively small, making it difficult to match the cascade with the input signal source. This problem disappears completely if you use a field-effect transistor - its input resistance reaches tens and even hundreds of megaohms. To get to know the field-effect transistor better, do the suggested experiments.

A little about the characteristics of a field-effect transistor. Like the bipolar one, the field electrode has three electrodes, but they are called differently: gate (similar to the base), drain (collector), source (emitter). By analogy with bipolar field-effect transistors, there are different “structures”: with a p-channel and an n-channel. Unlike bipolar ones, they can be with a gate in the form of a p-n junction and with an insulated gate. Our experiments will concern the first of them.

The basis of the field-effect transistor is a silicon wafer (gate), in which there is a thin region called a channel (Fig. 1a). On one side of the channel there is a drain, on the other there is a source. When connecting the positive terminal of the transistor to the source and the negative terminal of the power battery GB2 to the drain (Fig. 1, b), an electric current arises in the channel. The channel in this case has maximum conductivity.

As soon as you connect another power source - GB1 - to the source and gate terminals (plus to the gate), the channel “narrows”, causing an increase in resistance in the drain-source circuit. The current in this circuit immediately decreases. By changing the voltage between the gate and source, the drain current is controlled. Moreover, there is no current in the gate circuit; the drain current is controlled by an electric field (that’s why the transistor is called field effect), created by the voltage applied to the source and gate.

The above applies to a transistor with a p-channel, but if the transistor is with an n-channel, the polarity of the supply and control voltages is reversed (Fig. 1c).

Most often you can find a field-effect transistor in a metal case - then, in addition to the three main terminals, it may also have a housing terminal, which during installation is connected to the common wire of the structure.

One of the parameters of a field-effect transistor is the initial drain current (I from start), i.e., the current in the drain circuit at zero voltage at the transistor gate (in Fig. 2a, the variable resistor slider is in the lower position in the diagram) and at a given supply voltage .

If you smoothly move the resistor slider up in the circuit, then as the voltage at the transistor’s gate increases, the drain current decreases (Fig. 2b) and at a specific voltage for a given transistor it will drop to almost zero. The voltage corresponding to this moment is called the cut-off voltage (U ZIots).

The dependence of the drain current on the gate voltage is quite close to a straight line. If we take an arbitrary increment in the drain current and divide it by the corresponding increment in the voltage between the gate and source, we obtain the third parameter - the slope of the characteristic (S). This parameter is easy to determine without removing the characteristics or searching for it in the directory. It is enough to measure the initial drain current, and then connect, say, a galvanic element with a voltage of 1.5 V between the gate and the source. Subtract the resulting drain current from the initial one and divide the remainder by the element voltage - you get the value of the slope of the characteristic in milliamps per volt.

Knowledge of the features of a field-effect transistor will complement familiarity with its stock output characteristics (Fig. 2c). They are removed when the voltage between drain and source changes for several fixed gate voltages. It is easy to see that up to a certain voltage between the drain and the source, the output characteristic is nonlinear, and then within significant voltage limits it is almost horizontal.

Of course, a separate power supply is not used in real designs to supply bias voltage to the gate. The bias is formed automatically when a constant resistor of the required resistance is connected to the source circuit.

Now select several field-effect transistors of the KP103 (with p-channel), KP303 (with n-channel) series with different letter indices and practice determining their parameters using the given diagrams.

Field effect transistor is a touch sensor. The word "sensor" means feeling, sensation, perception. Therefore, we can assume that in our experiment the field-effect transistor will act as a sensitive element that reacts to touching one of its terminals.

In addition to the transistor (Fig. 3), for example, any of the KP103 series, you will need an ohmmeter with any measurement range. Connect the ohmmeter probes in any polarity to the drain and source terminals - the ohmmeter arrow will show a small resistance of this transistor circuit.

Then touch the shutter output with your finger. The ohmmeter needle will deviate sharply in the direction of increasing resistance. This happened because electrical current interference changed the voltage between the gate and source. The channel resistance increased, which was recorded by the ohmmeter.

Without removing your finger from the gate, try touching the source terminal with another finger. The ohmmeter needle will return to its original position - after all, the gate turned out to be connected through the resistance of the hand section to the source, which means that the control field between these electrodes has practically disappeared and the channel has become conductive.

These properties of field-effect transistors are often used in touch switches, buttons and switches.

Field effect transistor - field indicator. Modify the previous experiment slightly - bring the transistor with the gate terminal (or body) as close as possible to the power outlet or the wire of a working electrical appliance plugged into it. The effect will be the same as in the previous case - the ohmmeter needle will deviate in the direction of increasing resistance. This is understandable - an electric field is formed near the outlet or around the wire, to which the transistor reacts.

In this capacity, a field-effect transistor is used as a device sensor for detecting hidden electrical wiring or the location of a broken wire in a New Year's garland - at this point the field strength increases.

Holding the indicator transistor close to the power cord, try turning the electrical appliance on and off. The change in the electric field will be recorded by the ohmmeter needle.

Field effect transistor is a variable resistor. Having connected the bias voltage adjustment circuit between the gate and the source (Fig. 4), set the resistor slider to the bottom position according to the diagram. The ohmmeter needle, as in previous experiments, will record the minimum resistance of the drain-source circuit.

By moving the resistor slider up the circuit, you can observe a smooth change in the ohmmeter readings (increase in resistance). The field effect transistor has become a variable resistor with a very wide range of resistance changes, regardless of the value of the resistor in the gate circuit. The polarity of the ohmmeter connection does not matter, but the polarity of the galvanic element will have to be changed if a transistor with an n-channel is used, for example, any of the KP303 series. Field effect transistor - current stabilizer. To carry out this experiment (Fig. 5), you will need a direct current source with a voltage of 15...18 V (four series-connected 3336 batteries or an AC power supply), a variable resistor with a resistance of 10 or 15 kOhm, two constant resistors, a milliammeter with a measurement limit of 3- 5 mA, yes field effect transistor. First, set the resistor slider to the bottom position according to the diagram, corresponding to the supply of the minimum supply voltage to the transistor - about 5 V with the values ​​of resistors R2 and R3 indicated on the diagram. By selecting resistor R1 (if needed), set the current in the transistor drain circuit to 1.8...2.2 mA. As you move the resistor slider up the circuit, observe the change in drain current. It may happen that it will remain the same or increase slightly. In other words, when the supply voltage changes from 5 to 15...18 V, the current through the transistor will be automatically maintained at the specified level (by resistor R1). Moreover, the accuracy of current maintenance depends on the initially set value - the smaller it is, the higher the accuracy. Analysis of the stock output characteristics shown in Fig. will help confirm this conclusion. 2, c.

Such a cascade is called a current source or current generator. It can be found in a wide variety of designs.

Switching buck stabilizers

Y. SEMENOV, Rostov-on-Don

The article presented to our readers describes two pulsed step-down stabilizers: on discrete elements and on a specialized microcircuit. The first device is designed to power automotive equipment with a voltage of 12 Volts to the 24-volt on-board network of trucks and buses. The second device is the basis for a laboratory power supply.

Switching voltage stabilizers (step-down, step-up and inverting) occupy a special place in the history of the development of power electronics. Not so long ago, every power source with an output power of more than 50 W included a step-down switching stabilizer. Today, the scope of application of such devices has decreased due to the reduction in cost of power supplies with transformerless input. Nevertheless, the use of pulsed step-down stabilizers in some cases turns out to be more economically profitable than any other DC-voltage converters.

The functional diagram of a step-down switching stabilizer is shown in rice. 1, and timing diagrams explaining its operation in the mode of continuous inductor current L, ≈ on rice. 2. During t on, the electronic switch S is closed and the current flows through the circuit: positive terminal of the capacitor C in, resistive current sensor R dt, storage choke L, capacitor C out, load, negative terminal of the capacitor C in. At this stage, the inductor current l L is equal to the electronic commutator current S and increases almost linearly from l Lmin to l Lmax.

Based on a mismatch signal from the comparison node or an overload signal from a current sensor or a combination of both, the generator switches the electronic switch S to an open state. Since the current through the inductor L cannot change instantly, under the influence of self-induction emf the diode VD will open and the current l L will flow along the circuit: the cathode of the diode VD, the inductor L, the capacitor C Out, the load, the anode of the diode VD. At time t lKl, when the electronic commutator S is open, the inductor current l L coincides with the diode current VD and decreases linearly from

l Lmax to l L min . During Period T, capacitor C out receives and releases an increment of charge ΔQ out. corresponding to the shaded area on the time diagram of the current l L . This increment determines the range of ripple voltage ΔU Out on the capacitor C out and on the load.

When the electronic switch is closed, the diode closes. This process is accompanied by a sharp increase in the switch current to the value I smax due to the fact that the circuit resistance ≈ current sensor, closed switch, recovery diode ≈ is very small. To reduce dynamic losses, diodes with a short reverse recovery time should be used. In addition, the diodes of buck regulators must withstand high reverse current. With the restoration of the diode's closing properties, the next conversion period begins.

If a switching buck regulator operates at low load current, it may switch to intermittent inductor current mode. In this case, the inductor current stops at the moment the switch closes and its increase starts from zero. The intermittent current mode is undesirable when the load current is close to the rated current, since in this case increased output voltage ripple occurs. The most optimal situation is when the stabilizer operates in continuous inductor current mode at maximum load and in intermittent current mode when the load is reduced to 10...20% of the rated one.

The output voltage is regulated by changing the ratio of the time the switch is closed to the pulse repetition period. In this case, depending on the circuit design, various options for implementing the control method are possible. In devices with relay regulation, the transition from the on state of the switch to the off state is determined by the comparison node. When the output voltage is greater than the set voltage, the switch is turned off, and vice versa. If you fix the pulse repetition period, then the output voltage can be adjusted by changing the duration of the on state of the switch. Sometimes methods are used in which either the time of the closed or the time of the open state of the switch is recorded. In any of the control methods, it is necessary to limit the inductor current during the closed state of the switch to protect against output overload. For these purposes, a resistive sensor or pulse current transformer is used.

We will select the main elements of a pulse-step-down stabilizer and calculate their modes using a specific example. All relationships that are used in this case are obtained based on the analysis of the functional diagram and timing diagrams, and the methodology is taken as a basis.

1. Based on a comparison of the initial parameters and the maximum permissible values ​​of current and voltage of a number of powerful transistors and diodes, we first select the bipolar composite transistor KT853G (electronic switch S) and the diode KD2997V (VD).

2. Calculate the minimum and maximum fill factors:

γ min =t and min /T min =(U BуX +U pr)/(U BX max +U sincl ≈ U RдТ +U pr)=(12+0.8)/(32-2-0.3+ 0.8)=0.42;

γ max = t and max /T max = (U Bыx +U pp)/(U Bx min - U sbkl -U Rdt +U pp)=(12+0.8)/(18-2-0.3+ 0.8)=0.78, where U pp =0.8 V ≈ forward voltage drop across the diode VD, obtained from the forward branch of the I-V characteristic for a current equal to I Out in the worst case; U sbcl = 2 V ≈ saturation voltage of the KT853G transistor, performing the function of a switch S, with a current transfer coefficient in saturation mode h 21e = 250; U RдТ = 0.3 V ≈ voltage drop across the current sensor at rated load current.

3. Select the maximum and minimum conversion frequency.

This item is carried out if the pulse repetition period is not constant. We select a control method with a fixed duration of the open state of the electronic switch. In this case, the following condition is satisfied: t=(1 - γ max)/f min = (1 -γ min)/f max =const.

Since the switch is made on the KT853G transistor, which has poor dynamic characteristics, we will choose the maximum conversion frequency relatively low: f max = 25 kHz. Then the minimum conversion frequency can be defined as

f min =f max (1 - γ max)/(1 - γ min) =25╥10 3 ](1 - 0.78)/(1-0.42)=9.48 kHz.

4. Let's calculate the power loss on the switch.

Static losses are determined by the effective value of the current flowing through the switch. Since the current shape is ≈ trapezoidal, then I s = I out where α=l Lmax /l lx =1.25 ≈ the ratio of the maximum inductor current to the output current. Coefficient a is chosen within the range of 1.2... 1.6. Static losses of the switch P Scstat =l s U SBKn =3.27-2=6.54 W.

Dynamic losses on the switch Р sdin =0.5f max *U BX max (l smax *t f +α*l lx *t cn),

where I smax ≈ switch current amplitude due to reverse recovery of the diode VD. Taking l Smax =2l BуX , we obtain

Р sdin =0.5f max* U BX max * I out (2t f + α∙t cn)=0.5*25*10 3 *32*5(2*0.78-10 -6 +1.25 -2-10 -6)=8.12 W, where t f =0.78*10 -6 s ≈ duration of the front of the current pulse through the switch, t cn =2*10 -6 s ≈ decay duration.

The total losses on the switch are: Р s = Р sctat + Р sdin = 6.54 + 8.12 = 14.66 W.

If static losses were predominant on the switch, the calculation should have been carried out for the minimum input voltage when the inductor current is maximum. In cases where it is difficult to predict the prevailing type of losses, they are determined at both the minimum and maximum input voltage.

5. Calculate the power loss on the diode.

Since the shape of the current through the diode is also trapezoidal, we define its effective value as Static losses on the diode P vDcTaT =l vD ╥U pr =3.84-0.8=3.07 W.

The dynamic losses of the diode are mainly due to losses during reverse recovery: P VDdin =0.5f max *l smax *U Bx max *t oB *f max *l Bуx *U in max *t ov =25-10 3 -5-32 *0.2*10 -6 =0.8 W, where t OB =0.2-1C -6 s ≈ diode reverse recovery time.

The total losses on the diode will be: P VD =P MDstat +P VDdin =3.07+0.8=3.87 W.

6. Select a heat sink.

The main characteristic of a heat sink is its thermal resistance, which is defined as the ratio between the temperature difference between the environment and the surface of the heat sink to the power dissipated by it: R g =ΔТ/Р dissipation. In our case, the switching transistor and diode should be secured to the same heat sink through insulating spacers. In order not to take into account the thermal resistance of the gaskets and not to complicate the calculation, we choose a low surface temperature, approximately 70°C. Then at an ambient temperature of 40╟СΔТ=70-40=30╟С. The thermal resistance of the heat sink for our case is R t =ΔT/(P s +P vd)=30/(14.66+3.87)=1.62╟С/W.

Thermal resistance for natural cooling is usually given in the reference data for the heat sink. To reduce the size and weight of the device, you can use forced cooling using a fan.

7. Let's calculate the throttle parameters.

Let's calculate the inductance of the inductor:

L= (U BX max - U sbkl -U Rdt - U Out)γ min /=(32-2-0.3-12)*0.42/=118.94 µH.

As the material for the magnetic circuit, we choose MP 140 pressed with Mo-permalloy. The variable component of the magnetic field in the magnetic core in our case is such that hysteresis losses are not a limiting factor. Therefore, the maximum induction can be selected in the linear section of the magnetization curve near the inflection point. Working on a curved section is undesirable, since in this case the magnetic permeability of the material will be less than the initial one. This, in turn, will cause the inductance to decrease as the inductor current increases. We select the maximum induction B m equal to 0.5 T and calculate the volume of the magnetic circuit:

Vp=μμ 0 *L(αI vyx) 2 /B m 2 =140*4π*10 -7 *118.94* 10 -6 (1.25-5) 2 0.5 2 =3.27 cm 3, where μ=140 ≈

initial magnetic permeability of the material MP140; μ 0 =4π*10 -7 H/m ≈ magnetic constant.

Based on the calculated volume, we select the magnetic circuit. Due to the design features, the MP140 permalloy magnetic circuit is usually made on two folded rings. In our case, KP24x13x7 rings are suitable. The cross-sectional area of ​​the magnetic circuit is Sc = 20.352 = 0.7 cm 2, and the average length of the magnetic line is λc = 5.48 cm. The volume of the selected magnetic circuit is:

VC=SC* λс=0.7*5.48=3.86 cm 3 >Vp.

We calculate the number of turns: We take the number of turns equal to 23.

We will determine the diameter of the wire with insulation based on the fact that the winding must fit in one layer, turn to turn along the inner circumference of the magnetic circuit: d from =πd K k 3 /w=π*13-0.8/23= 1.42 mm, where d K =13 mm ≈ internal diameter of the magnetic circuit; k 3 =0.8 ≈ filling factor of the magnetic circuit window with the winding.

We choose PETV-2 wire with a diameter of 1.32 mm.

Before winding the wire, the magnetic circuit should be insulated with a PET-E film 20 microns thick and 6...7 mm wide in one layer.

8. Let's calculate the capacitance of the output capacitor: C Bуx =(U BX max -U sBkl - U Rдт) *γ min /=(32-2-0.3)*0.42/ =1250 μF, where ΔU Bуx =0, 01 V ≈ ripple range on the output capacitor.

The above formula does not take into account the influence of the internal, series resistance of the capacitor on ripple. Taking this into account, as well as a 20% tolerance for the capacitance of oxide capacitors, we select two K50-35 capacitors for a rated voltage of 40 V with a capacity of 1000 μF each. The choice of capacitors with an increased rated voltage is due to the fact that as this parameter increases, the series resistance of the capacitors decreases.

The diagram developed in accordance with the results obtained during the calculation is shown in rice. 3. Let's take a closer look at the operation of the stabilizer. During the open state of the electronic switch ≈ transistor VT5 ≈ a sawtooth voltage is formed on resistor R14 (current sensor). When it reaches a certain value, transistor VT3 will open, which, in turn, will open transistor VT2 and discharge capacitor S3. In this case, transistors VT1 and VT5 will close, and the switching diode VD3 will open. Previously open transistors VT3 and VT2 will close, but transistor VT1 will not open until the voltage on the capacitor SZ reaches the threshold level corresponding to its opening voltage. Thus, a time interval will be formed during which the switching transistor VT5 will be closed (approximately 30 μs). At the end of this interval, transistors VT1 and VT5 will open and the process will repeat again.

Resistor R. 10 and capacitor C4 form a filter that suppresses the voltage surge at the base of transistor VT3 due to reverse recovery of diode VD3.

For silicon transistor VT3, the base-emitter voltage at which it goes into active mode is about 0.6 V. In this case, relatively large power is dissipated at the current sensor R14. To reduce the voltage at the current sensor at which transistor VT3 opens, a constant bias of about 0.2 V is supplied to its base through the VD2R7R8R10 circuit.

A voltage proportional to the output voltage is supplied to the base of transistor VT4 from a divider, the upper arm of which is formed by resistors R15, R12, and the lower arm is formed by resistor R13. Circuit HL1R9 generates a reference voltage equal to the sum of the forward voltage drop across the LED and the emitter junction of transistor VT4. In our case, the reference voltage is 2.2 V. The mismatch signal is equal to the difference between the voltage at the base of transistor VT4 and the reference voltage.

The output voltage is stabilized by summing the mismatch signal amplified by transistor VT4 with the voltage based on transistor VT3. Let's assume that the output voltage has increased. Then the voltage at the base of transistor VT4 will become greater than the exemplary one. Transistor VT4 will open slightly and shift the voltage at the base of transistor VT3 so that it also begins to open. Consequently, transistor VT3 will open at a lower level of sawtooth voltage across resistor R14, which will lead to a reduction in the time interval at which the switching transistor will be open. The output voltage will then decrease.

If the output voltage decreases, the regulation process will be similar, but occurs in the reverse order and leads to an increase in the open time of the switch. Since the current of resistor R14 is directly involved in the formation of the open state time of transistor VT5, here, in addition to the usual output voltage feedback, there is current feedback. This allows you to stabilize the output voltage without load and ensure a quick response to sudden changes in current at the device output.

In the event of a short circuit in the load or overload, the stabilizer goes into current limiting mode. The output voltage begins to decrease at a current of 5.5...6 A, and the circuit current is approximately 8 A. In these modes, the on-state time of the switching transistor is reduced to a minimum, which reduces the power dissipated on it.

If the stabilizer malfunctions, caused by the failure of one of the elements (for example, breakdown of transistor VT5), the voltage at the output increases. In this case, the load may fail. To prevent emergency situations, the converter is equipped with a protection unit, which consists of a thyristor VS1, a zener diode VD1, a resistor R1 and a capacitor C1. When the output voltage exceeds the stabilization voltage of the zener diode VD1, a current begins to flow through it, which turns on the thyristor VS1. Its inclusion leads to a decrease in the output voltage to almost zero and the blowing of fuse FU1.

The device is designed to power 12-volt audio equipment, designed mainly for passenger vehicles, from the on-board network of trucks and buses with a voltage of 24 V. Due to the fact that the input voltage in this case has a low ripple level, capacitor C2 has a relatively small capacitance. It is insufficient when the stabilizer is powered directly from a mains transformer with a rectifier. In this case, the rectifier should be equipped with a capacitor with a capacity of at least 2200 μF for the corresponding voltage. The transformer must have an overall power of 80... 100 W.

The stabilizer uses oxide capacitors K50-35 (C2, C5, C6). Capacitor SZ ≈ film capacitor K73-9, K73-17, etc. of suitable sizes, C4 ≈ ceramic with low self-inductance, for example, K10-176. All resistors, except R14, ≈ C2-23 of the corresponding power. Resistor R14 is made from a 60 mm long piece of PEK 0.8 constantan wire with a linear resistance of approximately 1 Ohm/m.

A drawing of a printed circuit board made of one-sided foil fiberglass is shown in rice. 4.

Diode VD3, transistor VD5 and thyristor VS1 are attached to the heat sink through an insulating heat-conducting gasket using plastic bushings. The board is also attached to the same heat sink. The appearance of the assembled device is shown in rice. 5.

REFERENCES 1. Titze U., Schenk K. Semiconductor circuitry: A reference guide. Per. with him. ≈ M.: Mir, 1982. 2. Semiconductor devices. Medium and high power transistors: Handbook / A. A. Zaitsev, A. I. Mirkin, V. V. Mo-kryakov, etc. Ed. A. V. Golomedova. ≈ M.: Radio and communications, 1989. 3. Semiconductor devices. Rectifier diodes, zener diodes, thyristors: Handbook / A. B. Gitsevich, A. A. Zaitsev, V. V. Mokryakov, etc. Ed. A. V. Golomedova. ≈ M.: Radio and communication, 1988. 4 http:/ /www. ferrite.ru

Stabilized single-ended voltage converter

Radio magazine, number 3, 1999.

The article describes the principles of construction and a practical version of a simple pulse stabilized voltage converter that provides operation over a wide range of input voltage changes.

Among the various secondary power sources (SPS) with a transformerless input, the single-cycle self-oscillator converter with a “reverse” connection of the rectifier diode is distinguished by its extreme simplicity (Fig. 1).

Let us first briefly consider the operating principle of an unstabilized voltage converter, and then the method of stabilizing it.

Transformer T1 - linear choke; The intervals of energy accumulation in it and the transfer of accumulated energy to the load are spaced in time. In Fig. 2 shows: I I - current of the primary winding of the transformer, I II - current of the secondary winding, t n - interval of energy accumulation in the inductor, t p - interval of energy transfer to the load.

When the supply voltage U is connected, the base current of transistor VT1 begins to pass through resistor R1 (diode VD1 prevents the flow of current through the base winding circuit, and the capacitor C2 that shunts it increases positive feedback (POF) at the stage of forming voltage fronts). The transistor opens slightly, the PIC circuit closes through transformer T1, in which the regenerative process of energy storage occurs. Transistor VT1 enters saturation. The supply voltage is applied to the primary winding of the transformer, and the current I I (collector current I to transistor VT1) increases linearly. The base current I B of the saturated transistor is determined by the voltage on winding I II and the resistance of resistor R2. At the energy storage stage, diode VD2 is closed (hence the name of the converter - with “reverse” inclusion of the diode), and power consumption from the transformer occurs only by the input circuit of the transistor through the base winding.

When the collector current Ik reaches the value:

I K max = h 21E I B, (1)

where h 21E is the static current transfer coefficient of transistor VT1, the transistor leaves the saturation mode and a reverse regenerative process develops: the transistor closes, the diode VD2 opens and the energy accumulated by the transformer is transferred to the load. After the secondary winding current decreases, the energy storage stage begins again. The time interval t p is maximum when the converter is turned on, when the capacitor SZ is discharged and the voltage across the load is zero.

B shows that the power supply assembled according to the diagram in Fig. 1, - functional converter of the supply voltage source U power into the load current source I n.

It is important to note: since the stages of energy accumulation and transmission are separated in time, the maximum collector current of the transistor does not depend on the load current, i.e. the converter is completely protected from short circuits at the output. However, when the converter is turned on without a load (idle mode), a voltage surge on the transformer winding at the moment the transistor closes can exceed the maximum permissible value of the collector-emitter voltage and damage it.

The disadvantage of the simplest converter is the dependence of the collector current I K max, and therefore the output voltage, on the static current transfer coefficient of the transistor VT1. Therefore, the power supply parameters will vary significantly when using different instances.

A converter using a “self-protected” switching transistor has much more stable characteristics (Fig. 3).

A sawtooth voltage from resistor R3, proportional to the current of the primary winding of the transformer, is applied to the base of the auxiliary transistor VT2. As soon as the voltage across resistor R3 reaches the opening threshold of transistor VT2 (about 0.6 V), it will open and limit the base current of transistor VT1, which will interrupt the process of energy accumulation in the transformer. Maximum current of the primary winding of the transformer

I I max = I K max = 0.6/R3 (2)

turns out to be little dependent on the parameters of a particular transistor instance. Naturally, the current limit value calculated by formula (2) must be less than the current determined by formula (1) for the worst value of the static current transfer coefficient.

Now let's consider the possibility of regulating (stabilizing) the output voltage of the power supply.

B shows that the only parameter of the converter that can be changed to regulate the output voltage is the current I K max, or, what is the same, the energy accumulation time t n in the transformer, and the control (stabilization) unit can only reduce the current compared to the value , calculated according to formula (2).

Formulating the operating principle of the converter stabilization unit, the following requirements for it can be determined: - the constant output voltage of the converter must be compared with the reference voltage and, depending on their ratio, generate a mismatch voltage used to control the current I K max; - the process of current increase in the primary winding of the transformer should be controlled and stopped when it reaches a certain threshold determined by the mismatch voltage; - the control unit must provide galvanic isolation between the converter output and the switching transistor.

The control nodes implementing this algorithm shown in the diagrams contain a K521SAZ comparator, seven resistors, a transistor, a diode, two zener diodes and a transformer. Other well-known devices, including television power supplies, are also quite complex. Meanwhile, using a self-protected switching transistor, you can build a much simpler stabilized converter (see diagram in Fig. 4).

Feedback winding (OS) III and circuit VD3C4 form a feedback voltage proportional to the output voltage of the converter.

The reference stabilization voltage of the zener diode VD4 is subtracted from the feedback voltage, and the resulting mismatch signal is applied to resistor R5.

From the engine of trimming resistor R5, the sum of two voltages is supplied to the base of transistor VT2: a constant control voltage (part of the mismatch voltage) and a sawtooth voltage from resistor R3, proportional to the current of the primary winding of the transformer. Since the opening threshold of transistor VT2 is constant, an increase in the control voltage (for example, with an increase in the supply voltage U power and, accordingly, an increase in the output voltage of the converter) leads to a decrease in the current I I at which the transistor VT2 opens, and to a decrease in the output voltage. Thus, the converter becomes stabilized, and its output voltage is regulated within small limits by resistor R5.

The stabilization coefficient of the converter depends on the ratio of the change in the output voltage of the converter to the corresponding change in the constant voltage component based on transistor VT2. To increase the stabilization coefficient, it is necessary to increase the feedback voltage (the number of turns of winding III) and select the VD4 zener diode according to the stabilization voltage, which is less than the OS voltage by about 0.5 V. The widely used zener diodes of the D814 series with an OS voltage of about 10 V are practically quite suitable.

It should be noted that to achieve better temperature stability of the converter, it is necessary to use a zener diode VD4 with a positive TKN, which compensates for the decrease in voltage drop across the emitter junction of transistor VT2 when heated. Therefore, the D814 series zener diodes are more suitable than the D818 precision zener diodes.

The number of output windings of the transformer (similar to winding II) can be increased, i.e. the converter can be made multi-channel.

Built according to the diagram in Fig. 4 converters provide good stabilization of output voltages when the input voltage changes within a very wide range (150...250 V). However, when operating on a variable load, especially in multi-channel converters, the results are somewhat worse, since when the load current changes in one of the windings, energy is redistributed between all windings. In this case, the change in the feedback voltage reflects the change in the output voltage of the converter with less accuracy.

It is possible to improve stabilization when operating on a variable load if the OS voltage is generated directly from the output voltage. The easiest way to do this is to use an additional low-power transformer voltage converter assembled according to any of the known circuits.

The use of an additional voltage converter is also justified in the case of a multi-channel power source. The high-voltage converter provides one of the stabilized voltages (the highest of them - at high voltages, the capacitor filter at the output of the converter is more efficient), and the remaining voltages, including the OS voltage, are generated by an additional converter.

For the manufacture of a transformer, it is best to use an armored ferrite magnetic core with a gap in the central rod, which ensures linear magnetization. If there is no such magnetic circuit, you can use a 0.1...0.3 mm thick gasket made of PCB or even paper to create a gap. It is also possible to use ring magnetic cores.

Although the literature indicates that for the converters with a “reverse” diode connection considered in this article, the output filter can be purely capacitive, the use of LC filters can further reduce the output voltage ripple.

For safe operation of the IVEP, a trimming resistor (R5 in Fig. 4) with good insulation of the engine should be used. The transformer windings, galvanically connected to the mains voltage, must be reliably insulated from the output. The same applies to other radioelements.

Like any power source with frequency conversion, the described power source must be equipped with an electromagnetic shield and an input filter.

The safety of setting up the converter will be ensured by a network transformer with a transformation ratio equal to unity. However, it is best to use a series-connected LATR and an isolation transformer.

Turning on the converter without a load will most likely lead to breakdown of the powerful switching transistor. Therefore, before you start setting up, connect the equivalent load. After switching on, you should first check the voltage on resistor R3 with an oscilloscope - it should increase linearly at stage t n. If linearity is broken, this means that the magnetic circuit is entering saturation and the transformer must be recalculated. Using a high-voltage probe, check the signal at the collector of the switching transistor - the pulse declines should be quite steep, and the voltage on the open transistor should be small. If necessary, you should adjust the number of turns of the base winding and the resistance of resistor R2 in the transistor base circuit.

Next, you can try to change the output voltage of the converter with resistor R5; if necessary, adjust the number of turns of the OS winding and select a VD4 zener diode. Check the operation of the converter when the input voltage and load change.

In Fig. Figure 5 shows an IVEP diagram for a ROM programmer as an example of using a converter built on the basis of the proposed principle.

Source parameters are given in table. 1.

When the mains voltage changes from 140 to 240 V, the voltage at the output of the 28 V source is within the range of 27.6...28.2 V; source +5 V - 4.88...5 V.

Capacitors C1-SZ and inductor L1 form an input mains filter that reduces the emission of high-frequency interference by the converter. Resistor R1 limits the charging current pulse of capacitor C4 when the converter is turned on.

Circuit R3C5 smoothes out voltage surges on transistor VT1 (a similar circuit is not shown in the previous figures).

A conventional converter is assembled on transistors VT3, VT4, generating two more from the output voltage +28 V: +5 V and -5 V, as well as the OS voltage. In general, the IVEP provides a stabilized voltage of +28 V. The stability of the other two output voltages is ensured by powering an additional converter from a +28 V source and a fairly constant load on these channels.

The IVEP provides protection against exceeding the output voltage of +28 V to 29 V. When exceeded, the triac VS1 opens and closes the +28 V source. The power supply emits a loud squeak. The current through the triac is 0.75 A.

Transistor VT1 is installed on a small heat sink made of an aluminum plate measuring 40 (30 mm). Instead of the KT828A transistor, you can use other high-voltage devices with a voltage of at least 600 V and a current of more than 1 A, for example, KT826B, KT828B, KT838A.

Instead of the KT3102A transistor, you can use any KT3102 series; transistors KT815G can be replaced with KT815V, KT817V, KT817G. Rectifier diodes (except VD1) must be used with high frequencies, for example, the KD213 series, etc. It is advisable to use oxide filter capacitors of the K52, ETO series. Capacitor C5 must have a voltage of at least 600 V.

The TS106-10 (VS1) triac is used solely because of its small size. Almost any type of SCR that can withstand a current of about 1 A is suitable, including the KU201 series. However, the thyristor will have to be selected according to the minimum control current.

It should be noted that in a particular case (with relatively small current consumption from the source) it would be possible to do without a second converter by building a converter according to the circuit in Fig. 4 with additional windings for +5 V and -5 V channels and linear stabilizers of the KR142 series. The use of an additional converter is caused by the desire to conduct comparative studies of various IVEPs and make sure that the proposed option provides better output voltage stabilization.

The parameters of transformers and chokes are given in table. 2.

Table 2
Designation	Magnetic core		Number of turns
	B26 M1000 with a gap in the central rod			PEV-2 0.18 PEV-2 0.35 PEV-2 0.18
	K16x10x4.5 M2000NM1		2x65 2x7 2x13 23	PEV-2 0.18 PEV-2 0.18 PEV-2 0.35 MGTF 0.07
	K16x10x4.5 M2000NM1	MGTF 0.07 in two wires until filled
	K17.5x8x5 M2000NM1
	K16x10x4.5 M2000NM1
	K12x5x5.5 M2000NM1

The magnetic core for transformer T1 is used from the filter choke of the power supply of the drive on removable magnetic disks of the ES series of computers.

The types of magnetic circuits of chokes L1-L4 are not critical.

The source is set up according to the above method, but first the overvoltage protection should be turned off by moving the resistor R10 slider to the bottom position according to the diagram. After setting up the IVEP, you should use resistor R5 to set the output voltage to +29 V and, slowly rotating the slider of resistor R10, reach the opening threshold of triac VS1. Then turn off the source, turn the slider of resistor R5 towards decreasing the output voltage, turn on the source and use resistor R5 to set the output voltage to 28 V.

It should be noted: since the voltages at the +5 V and -5 V outputs depend on the +28 V voltage and are not regulated separately from it, depending on the parameters of the elements used and the current of a particular load, it may be necessary to select the number of turns of the windings of the T2 transformer.

Literature

1. Bas A. A., Milovzorov V. P., Musolin A. K. Secondary power supplies with transformerless input. - M.: Radio and communication, 1987.

Making a power supply with your own hands makes sense not only for enthusiastic radio amateurs. A homemade power supply unit (PSU) will create convenience and save a considerable amount in the following cases:

• To power low-voltage power tools, to save the life of an expensive rechargeable battery;
• For electrification of premises that are particularly dangerous in terms of the degree of electric shock: basements, garages, sheds, etc. When powered by alternating current, a large amount of it in low-voltage wiring can create interference with household appliances and electronics;
• In design and creativity for precise, safe and waste-free cutting of foam plastic, foam rubber, low-melting plastics with heated nichrome;
• In lighting design, the use of special power supplies will extend the life of the LED strip and obtain stable lighting effects. Powering underwater illuminators, etc. from a household electrical network is generally unacceptable;
• For charging phones, smartphones, tablets, laptops away from stable power sources;
• For electroacupuncture;
• And many other purposes not directly related to electronics.

Acceptable simplifications

Professional power supplies are designed to power any kind of load, incl. reactive. Possible consumers include precision equipment. The pro-BP must maintain the specified voltage with the highest accuracy for an indefinitely long time, and its design, protection and automation must allow operation by unqualified personnel in difficult conditions, for example. biologists to power their instruments in a greenhouse or on an expedition.

An amateur laboratory power supply is free from these limitations and therefore can be significantly simplified while maintaining quality indicators sufficient for personal use. Further, through also simple improvements, it is possible to obtain a special-purpose power supply from it. What are we going to do now?

Abbreviations

1. KZ – short circuit.
2. XX – idle speed, i.e. sudden disconnection of the load (consumer) or a break in its circuit.
3. VS – voltage stabilization coefficient. It is equal to the ratio of the change in input voltage (in % or times) to the same output voltage at a constant current consumption. Eg. The network voltage dropped completely, from 245 to 185V. Relative to the norm of 220V, this will be 27%. If the VS of the power supply is 100, the output voltage will change by 0.27%, which, with its value of 12V, will give a drift of 0.033V. More than acceptable for amateur practice.
4. IPN is a source of unstabilized primary voltage. This can be an iron transformer with a rectifier or a pulsed network voltage inverter (VIN).
5. IIN - operate at a higher (8-100 kHz) frequency, which allows the use of lightweight compact ferrite transformers with windings of several to several dozen turns, but they are not without drawbacks, see below.
6. RE – regulating element of the voltage stabilizer (SV). Maintains the output at its specified value.
7. ION – reference voltage source. Sets its reference value, according to which, together with the OS feedback signals, the control device of the control unit influences the RE.
8. SNN – continuous voltage stabilizer; simply “analog”.
9. ISN – pulse voltage stabilizer.
10. UPS is a switching power supply.

Note: both SNN and ISN can operate both from an industrial frequency power supply with a transformer on iron, and from an electrical power supply.

About computer power supplies

UPSs are compact and economical. And in the pantry many people have a power supply from an old computer lying around, obsolete, but quite serviceable. So is it possible to adapt a switching power supply from a computer for amateur/working purposes? Unfortunately, a computer UPS is a rather highly specialized device and the possibilities of its use at home/at work are very limited:

It is perhaps advisable for the average amateur to use a UPS converted from a computer one only to power power tools; about this see below. The second case is if an amateur is engaged in PC repair and/or creation of logic circuits. But then he already knows how to adapt a power supply from a computer for this:

1. Load the main channels +5V and +12V (red and yellow wires) with nichrome spirals at 10-15% of the rated load;
2. The green soft start wire (low-voltage button on the front panel of the system unit) pc on is shorted to common, i.e. on any of the black wires;
3. On/off is performed mechanically, using a toggle switch on the rear panel of the power supply unit;
4. With mechanical (iron) I/O “on duty”, i.e. independent power supply of USB ports +5V will also be turned off.

Get to work!

Due to the shortcomings of UPSs, plus their fundamental and circuitry complexity, we will only look at a couple of them at the end, but simple and useful, and talk about the method of repairing the IPS. The main part of the material is devoted to SNN and IPN with industrial frequency transformers. They allow a person who has just picked up a soldering iron to build a power supply of very high quality. And having it on the farm, it will be easier to master “fine” techniques.

IPN

First, let's look at the IPN. We’ll leave pulse ones in more detail until the section on repairs, but they have something in common with “iron” ones: a power transformer, a rectifier and a ripple suppression filter. Together, they can be implemented in various ways depending on the purpose of the power supply.

Pos. 1 in Fig. 1 – half-wave (1P) rectifier. The voltage drop across the diode is the smallest, approx. 2B. But the pulsation of the rectified voltage is with a frequency of 50 Hz and is “ragged”, i.e. with intervals between pulses, so the pulsation filter capacitor Sf should be 4-6 times larger in capacity than in other circuits. The use of power transformer Tr for power is 50%, because Only 1 half-wave is rectified. For the same reason, a magnetic flux imbalance occurs in the Tr magnetic circuit and the network “sees” it not as an active load, but as inductance. Therefore, 1P rectifiers are used only for low power and where there is no other way, for example. in IIN on blocking generators and with a damper diode, see below.

Note: why 2V, and not 0.7V, at which the p-n junction in silicon opens? The reason is through current, which is discussed below.

Pos. 2 – 2-half-wave with midpoint (2PS). The diode losses are the same as before. case. The ripple is 100 Hz continuous, so the smallest possible Sf is needed. Use of Tr - 100% Disadvantage - double copper consumption on the secondary winding. At the time when rectifiers were made using kenotron lamps, this did not matter, but now it is decisive. Therefore, 2PS are used in low-voltage rectifiers, mainly at higher frequencies with Schottky diodes in UPSs, but 2PS have no fundamental limitations on power.

Pos. 3 – 2-half-wave bridge, 2RM. Losses on diodes are doubled compared to pos. 1 and 2. The rest is the same as 2PS, but the secondary copper is needed almost half as much. Almost - because several turns have to be wound to compensate for the losses on a pair of “extra” diodes. The most commonly used circuit is for voltages from 12V.

Pos. 3 – bipolar. The “bridge” is depicted conventionally, as is customary in circuit diagrams (get used to it!), and is rotated 90 degrees counterclockwise, but in fact it is a pair of 2PS connected in opposite polarities, as can be clearly seen further in Fig. 6. Copper consumption is the same as 2PS, diode losses are the same as 2PM, the rest is the same as both. It is built mainly to power analog devices that require voltage symmetry: Hi-Fi UMZCH, DAC/ADC, etc.

Pos. 4 – bipolar according to the parallel doubling scheme. Provides increased voltage symmetry without additional measures, because asymmetry of the secondary winding is excluded. Using Tr 100%, ripples 100 Hz, but torn, so Sf needs double capacity. Losses on the diodes are approximately 2.7V due to the mutual exchange of through currents, see below, and at a power of more than 15-20 W they increase sharply. They are built mainly as low-power auxiliary ones for independent power supply of operational amplifiers (op-amps) and other low-power, but demanding analog components in terms of power supply quality.

How to choose a transformer?

In a UPS, the entire circuit is most often clearly tied to the standard size (more precisely, to the volume and cross-sectional area Sc) of the transformer/transformers, because the use of fine processes in ferrite makes it possible to simplify the circuit while making it more reliable. Here, “somehow in your own way” comes down to strict adherence to the developer’s recommendations.

The iron-based transformer is selected taking into account the characteristics of the SNN, or is taken into account when calculating it. The voltage drop across the RE Ure should not be taken less than 3V, otherwise the VS will drop sharply. As Ure increases, the VS increases slightly, but the dissipated RE power grows much faster. Therefore, Ure is taken at 4-6 V. To it we add 2(4) V of losses on the diodes and the voltage drop on the secondary winding Tr U2; for a power range of 30-100 W and voltages of 12-60 V, we take it to 2.5 V. U2 arises primarily not from the ohmic resistance of the winding (it is generally negligible in powerful transformers), but due to losses due to magnetization reversal of the core and the creation of a stray field. Simply, part of the network energy, “pumped” by the primary winding into the magnetic circuit, evaporates into outer space, which is what the value of U2 takes into account.

So, we calculated, for example, for a bridge rectifier, 4 + 4 + 2.5 = 10.5 V extra. We add it to the required output voltage of the power supply unit; let it be 12V, and divide by 1.414, we get 22.5/1.414 = 15.9 or 16V, this will be the lowest permissible voltage of the secondary winding. If TP is factory-made, we take 18V from the standard range.

Now the secondary current comes into play, which, naturally, is equal to the maximum load current. Let us say we need 3A; multiply by 18V, it will be 54W. We have obtained the overall power Tr, Pg, and we will find the rated power P by dividing Pg by the efficiency Tr η, which depends on Pg:

• up to 10W, η = 0.6.
• 10-20 W, η = 0.7.
• 20-40 W, η = 0.75.
• 40-60 W, η = 0.8.
• 60-80 W, η = 0.85.
• 80-120 W, η = 0.9.
• from 120 W, η = 0.95.

In our case, there will be P = 54/0.8 = 67.5 W, but there is no such standard value, so you will have to take 80 W. In order to get 12Vx3A = 36W at the output. A steam locomotive, and that's all. It’s time to learn how to calculate and wind the “trances” yourself. Moreover, in the USSR, methods for calculating transformers on iron were developed that make it possible, without loss of reliability, to squeeze 600 W out of a core, which, when calculated according to amateur radio reference books, is capable of producing only 250 W. "Iron Trance" is not as stupid as it seems.

SNN

The rectified voltage needs to be stabilized and, most often, regulated. If the load is more powerful than 30-40 W, short-circuit protection is also necessary, otherwise a malfunction of the power supply may cause a network failure. SNN does all this together.

Simple reference

It is better for a beginner not to immediately go into high power, but to make a simple, highly stable 12V ELV for testing according to the circuit in Fig. 2. It can then be used as a source of reference voltage (its exact value is set by R5), for checking devices, or as a high-quality ELV ION. The maximum load current of this circuit is only 40mA, but the VSC on the antediluvian GT403 and the equally ancient K140UD1 is more than 1000, and when replacing VT1 with a medium-power silicon one and DA1 on any of the modern op-amps it will exceed 2000 and even 2500. The load current will also increase to 150 -200 mA, which is already useful.

0-30

The next stage is a power supply with voltage regulation. The previous one was done according to the so-called. compensating comparison circuit, but it is difficult to convert one to a high current. We will make a new SNN based on an emitter follower (EF), in which the RE and CU are combined in just one transistor. The KSN will be somewhere around 80-150, but this will be enough for an amateur. But the SNN on the ED allows, without any special tricks, to obtain an output current of up to 10A or more, as much as the Tr will give and the RE will withstand.

The circuit of a simple 0-30V power supply is shown in pos. 1 Fig. 3. IPN for it is a ready-made transformer such as TPP or TS for 40-60 W with a secondary winding for 2x24V. Rectifier type 2PS with diodes rated at 3-5A or more (KD202, KD213, D242, etc.). VT1 is installed on a radiator with an area of ​​50 square meters or more. cm; An old PC processor will work very well. Under such conditions, this ELV is not afraid of a short circuit, only VT1 and Tr will heat up, so a 0.5A fuse in the primary winding circuit of Tr is enough for protection.

Pos. Figure 2 shows how convenient a power supply on an electric power supply is for an amateur: there is a 5A power supply circuit with adjustment from 12 to 36 V. This power supply can supply 10A to the load if there is a 400W 36V power supply. Its first feature is the integrated SNN K142EN8 (preferably with index B) acts in an unusual role as a control unit: to its own 12V output is added, partially or completely, all 24V, the voltage from the ION to R1, R2, VD5, VD6. Capacitors C2 and C3 prevent excitation on HF DA1 operating in an unusual mode.

The next point is the short circuit protection device (PD) on R3, VT2, R4. If the voltage drop across R4 exceeds approximately 0.7V, VT2 will open, close the base circuit of VT1 to the common wire, it will close and disconnect the load from the voltage. R3 is needed so that the extra current does not damage DA1 when the ultrasound is triggered. There is no need to increase its denomination, because when the ultrasound is triggered, you need to securely lock VT1.

And the last thing is the seemingly excessive capacitance of the output filter capacitor C4. In this case it is safe, because The maximum collector current of VT1 of 25A ensures its charge when turned on. But this ELV can supply a current of up to 30A to the load within 50-70 ms, so this simple power supply is suitable for powering low-voltage power tools: its starting current does not exceed this value. You just need to make (at least from plexiglass) a contact block-shoe with a cable, put on the heel of the handle, and let the “Akumych” rest and save resources before leaving.

About cooling

Let's say in this circuit the output is 12V with a maximum of 5A. This is just the average power of a jigsaw, but, unlike a drill or screwdriver, it takes it all the time. At C1 it stays at about 45V, i.e. on RE VT1 it remains somewhere around 33V at a current of 5A. Power dissipation is more than 150 W, even more than 160, if you consider that VD1-VD4 also needs to be cooled. It is clear from this that any powerful adjustable power supply must be equipped with a very effective cooling system.

A finned/needle radiator using natural convection does not solve the problem: calculations show that a dissipating surface of 2000 sq. m. is needed. see and the thickness of the radiator body (the plate from which the fins or needles extend) is from 16 mm. To own this much aluminum in a shaped product was and remains a dream in a crystal castle for an amateur. A CPU cooler with airflow is also not suitable; it is designed for less power.

One of the options for the home craftsman is an aluminum plate with a thickness of 6 mm and dimensions of 150x250 mm with holes of increasing diameter drilled along the radii from the installation site of the cooled element in a checkerboard pattern. It will also serve as the rear wall of the power supply housing, as in Fig. 4.

An indispensable condition for the effectiveness of such a cooler is a weak, but continuous flow of air through the perforations from the outside to the inside. To do this, install a low-power exhaust fan in the housing (preferably at the top). A computer with a diameter of 76 mm or more is suitable, for example. add. HDD cooler or video card. It is connected to pins 2 and 8 of DA1, there is always 12V.

Note: In fact, a radical way to overcome this problem is a secondary winding Tr with taps for 18, 27 and 36V. The primary voltage is switched depending on which tool is being used.

And yet the UPS

The described power supply for the workshop is good and very reliable, but it’s hard to carry it with you on trips. This is where a computer power supply will fit in: the power tool is insensitive to most of its shortcomings. Some modification most often comes down to installing an output (closest to the load) electrolytic capacitor of large capacity for the purpose described above. There are a lot of recipes for converting computer power supplies for power tools (mainly screwdrivers, which are not very powerful, but very useful) in RuNet; one of the methods is shown in the video below, for a 12V tool.

Video: 12V power supply from a computer

With 18V tools it’s even easier: for the same power they consume less current. A much more affordable ignition device (ballast) from a 40 W or more energy saving lamp may be useful here; it can be completely placed in the case of a bad battery, and only the cable with the power plug will remain outside. How to make a power supply for an 18V screwdriver from ballast from a burnt housekeeper, see the following video.

Video: 18V power supply for a screwdriver

High class

But let’s return to SNN on ES; their capabilities are far from being exhausted. In Fig. 5 – bipolar powerful power supply with 0-30 V regulation, suitable for Hi-Fi audio equipment and other fastidious consumers. The output voltage is set using one knob (R8), and the symmetry of the channels is maintained automatically at any voltage value and any load current. A pedant-formalist may turn gray before his eyes when he sees this circuit, but the author has had such a power supply working properly for about 30 years.

The main stumbling block during its creation was δr = δu/δi, where δu and δi are small instantaneous increments of voltage and current, respectively. To develop and set up high-quality equipment, it is necessary that δr does not exceed 0.05-0.07 Ohm. Simply, δr determines the ability of the power supply to instantly respond to surges in current consumption.

For the SNN on the EP, δr is equal to that of the ION, i.e. zener diode divided by the current transfer coefficient β RE. But for powerful transistors, β drops significantly at a large collector current, and δr of a zener diode ranges from a few to tens of ohms. Here, in order to compensate for the voltage drop across the RE and reduce the temperature drift of the output voltage, we had to assemble a whole chain of them in half with diodes: VD8-VD10. Therefore, the reference voltage from the ION is removed through an additional ED on VT1, its β is multiplied by β RE.

The next feature of this design is short circuit protection. The simplest one, described above, does not fit into a bipolar circuit in any way, so the protection problem is solved according to the principle “there is no trick against scrap”: there is no protective module as such, but there is redundancy in the parameters of powerful elements - KT825 and KT827 at 25A and KD2997A at 30A. T2 is not capable of providing such a current, and while it warms up, FU1 and/or FU2 will have time to burn out.

Note: It is not necessary to indicate blown fuses on miniature incandescent lamps. It’s just that at that time LEDs were still quite scarce, and there were several handfuls of SMOKs in the stash.

It remains to protect the RE from the extra discharge currents of the pulsation filter C3, C4 during a short circuit. To do this, they are connected through low-resistance limiting resistors. In this case, pulsations may appear in the circuit with a period equal to the time constant R(3,4)C(3,4). They are prevented by C5, C6 of smaller capacity. Their extra currents are no longer dangerous for RE: the charge drains faster than the crystals of the powerful KT825/827 heat up.

Output symmetry is ensured by op-amp DA1. The RE of the negative channel VT2 is opened by current through R6. As soon as the minus of the output exceeds the plus in absolute value, it will slightly open VT3, which will close VT2 and the absolute values ​​of the output voltages will be equal. Operational control over the symmetry of the output is carried out using a dial gauge with a zero in the middle of the scale P1 (its appearance is shown in the inset), and adjustment, if necessary, is carried out by R11.

The last highlight is the output filter C9-C12, L1, L2. This design is necessary to absorb possible HF interference from the load, so as not to rack your brain: the prototype is buggy or the power supply is “wobbly”. With electrolytic capacitors alone, shunted with ceramics, there is no complete certainty here; the large self-inductance of the “electrolytes” interferes. And chokes L1, L2 divide the “return” of the load across the spectrum, and to each their own.

This power supply unit, unlike the previous ones, requires some adjustment:

1. Connect a load of 1-2 A at 30V;
2. R8 is set to maximum, in the highest position according to the diagram;
3. Using a reference voltmeter (any digital multimeter will do now) and R11, the channel voltages are set to be equal in absolute value. Maybe, if the op-amp does not have the ability to balance, you will have to select R10 or R12;
4. Use the R14 trimmer to set P1 exactly to zero.

About power supply repair

PSUs fail more often than other electronic devices: they take the first blow of network surges, and they also get a lot from the load. Even if you do not intend to make your own power supply, a UPS can be found, in addition to a computer, in a microwave oven, washing machine, and other household appliances. The ability to diagnose a power supply and knowledge of the basics of electrical safety will make it possible, if not to fix the fault yourself, then to competently bargain on the price with repairmen. Therefore, let's look at how a power supply is diagnosed and repaired, especially with an IIN, because over 80% of failures are their share.

Saturation and draft

First of all, about some effects, without understanding which it is impossible to work with a UPS. The first of them is the saturation of ferromagnets. They are not capable of absorbing energies of more than a certain value, depending on the properties of the material. Hobbyists rarely encounter saturation on iron; it can be magnetized to several Tesla (Tesla, a unit of measurement of magnetic induction). When calculating iron transformers, the induction is taken to be 0.7-1.7 Tesla. Ferrites can withstand only 0.15-0.35 T, their hysteresis loop is “more rectangular”, and operate at higher frequencies, so their probability of “jumping into saturation” is orders of magnitude higher.

If the magnetic circuit is saturated, the induction in it no longer grows and the EMF of the secondary windings disappears, even if the primary has already melted (remember school physics?). Now turn off the primary current. The magnetic field in soft magnetic materials (hard magnetic materials are permanent magnets) cannot exist stationary, like an electric charge or water in a tank. It will begin to dissipate, the induction will drop, and an EMF of the opposite polarity relative to the original polarity will be induced in all windings. This effect is quite widely used in IIN.

Unlike saturation, through current in semiconductor devices (simply draft) is an absolutely harmful phenomenon. It arises due to the formation/resorption of space charges in the p and n regions; for bipolar transistors - mainly in the base. Field-effect transistors and Schottky diodes are practically free from drafts.

For example, when voltage is applied/removed to a diode, it conducts current in both directions until the charges are collected/dissolved. That is why the voltage loss on the diodes in rectifiers is more than 0.7V: at the moment of switching, part of the charge of the filter capacitor has time to flow through the winding. In a parallel doubling rectifier, the draft flows through both diodes at once.

A draft of transistors causes a voltage surge on the collector, which can damage the device or, if a load is connected, damage it through extra current. But even without that, a transistor draft increases dynamic energy losses, like a diode draft, and reduces the efficiency of the device. Powerful field-effect transistors are almost not susceptible to it, because do not accumulate charge in the base due to its absence, and therefore switch very quickly and smoothly. “Almost”, because their source-gate circuits are protected from reverse voltage by Schottky diodes, which are slightly, but through.

TIN types

UPS trace their origins to the blocking generator, pos. 1 in Fig. 6. When turned on, Uin VT1 is slightly opened by current through Rb, current flows through winding Wk. It cannot instantly grow to the limit (remember school physics again); an emf is induced in the base Wb and load winding Wn. From Wb, through Sb, it forces the unlocking of VT1. No current flows through Wn yet and VD1 does not start up.

When the magnetic circuit is saturated, the currents in Wb and Wn stop. Then, due to the dissipation (resorption) of energy, the induction drops, an EMF of the opposite polarity is induced in the windings, and the reverse voltage Wb instantly locks (blocks) VT1, saving it from overheating and thermal breakdown. Therefore, such a scheme is called a blocking generator, or simply blocking. Rk and Sk cut off HF interference, of which blocking produces more than enough. Now some useful power can be removed from Wn, but only through the 1P rectifier. This phase continues until the Sat is completely recharged or until the stored magnetic energy is exhausted.

This power, however, is small, up to 10W. If you try to take more, VT1 will burn out from a strong draft before it locks. Since Tp is saturated, the blocking efficiency is no good: more than half of the energy stored in the magnetic circuit flies away to warm other worlds. True, due to the same saturation, blocking to some extent stabilizes the duration and amplitude of its pulses, and its circuit is very simple. Therefore, blocking-based TINs are often used in cheap phone chargers.

Note: the value of Sb largely, but not completely, as they write in amateur reference books, determines the pulse repetition period. The value of its capacitance must be linked to the properties and dimensions of the magnetic circuit and the speed of the transistor.

Blocking at one time gave rise to line scan TVs with cathode ray tubes (CRT), and it gave birth to an INN with a damper diode, pos. 2. Here the control unit, based on signals from Wb and the DSP feedback circuit, forcibly opens/locks VT1 before Tr is saturated. When VT1 is locked, the reverse current Wk is closed through the same damper diode VD1. This is the working phase: already greater than in blocking, part of the energy is removed into the load. It’s big because when it’s completely saturated, all the extra energy flies away, but here there’s not enough of that extra. In this way it is possible to remove power up to several tens of watts. However, since the control device cannot operate until Tr has approached saturation, the transistor still shows through strongly, the dynamic losses are large and the efficiency of the circuit leaves much more to be desired.

The IIN with a damper is still alive in televisions and CRT displays, since in them the IIN and the horizontal scan output are combined: the power transistor and TP are common. This greatly reduces production costs. But, frankly speaking, an IIN with a damper is fundamentally stunted: the transistor and transformer are forced to work all the time on the verge of failure. The engineers who managed to bring this circuit to acceptable reliability deserve the deepest respect, but it is strongly not recommended to stick a soldering iron in there except for professionals who have undergone professional training and have the appropriate experience.

The push-pull INN with a separate feedback transformer is most widely used, because has the best quality indicators and reliability. However, in terms of RF interference, it also sins terribly in comparison with “analog” power supplies (with transformers on hardware and SNN). Currently, this scheme exists in many modifications; powerful bipolar transistors in it are almost completely replaced by field-effect ones controlled by special devices. IC, but the principle of operation remains unchanged. It is illustrated by the original diagram, pos. 3.

The limiting device (LD) limits the charging current of the capacitors of the input filter Sfvkh1(2). Their large size is an indispensable condition for the operation of the device, because During one operating cycle, a small fraction of the stored energy is taken from them. Roughly speaking, they play the role of a water tank or air receiver. When charging “short”, the extra charge current can exceed 100A for a time of up to 100 ms. Rc1 and Rc2 with a resistance of the order of MOhm are needed to balance the filter voltage, because the slightest imbalance of his shoulders is unacceptable.

When Sfvkh1(2) are charged, the ultrasonic trigger device generates a trigger pulse that opens one of the arms (which one does not matter) of the inverter VT1 VT2. A current flows through the winding Wk of a large power transformer Tr2 and the magnetic energy from its core through the winding Wn is almost completely spent on rectification and on the load.

A small part of the energy Tr2, determined by the value of Rogr, is removed from the winding Woc1 and supplied to the winding Woc2 of a small basic feedback transformer Tr1. It quickly saturates, the open arm closes and, due to dissipation in Tr2, the previously closed one opens, as described for blocking, and the cycle repeats.

In essence, a push-pull IIN is 2 blockers “pushing” each other. Since the powerful Tr2 is not saturated, the draft VT1 VT2 is small, completely “sinks” into the magnetic circuit Tr2 and ultimately goes into the load. Therefore, a two-stroke IPP can be built with a power of up to several kW.

It's worse if he ends up in XX mode. Then, during the half cycle, Tr2 will have time to saturate itself and a strong draft will burn both VT1 and VT2 at once. However, now there are power ferrites on sale for induction up to 0.6 Tesla, but they are expensive and degrade from accidental magnetization reversal. Ferrites with a capacity of more than 1 Tesla are being developed, but in order for IINs to achieve “iron” reliability, at least 2.5 Tesla is needed.

Diagnostic technique

When troubleshooting an “analog” power supply, if it is “stupidly silent,” first check the fuses, then the protection, RE and ION, if it has transistors. They ring normally - we move on element by element, as described below.

In the IIN, if it “starts up” and immediately “stalls out”, they first check the control unit. The current in it is limited by a powerful low-resistance resistor, then shunted by an optothyristor. If the “resistor” is apparently burnt, replace it and the optocoupler. Other elements of the control device fail extremely rarely.

If the IIN is “silent, like a fish on ice,” the diagnosis also begins with the OU (maybe the “rezik” has completely burned out). Then - ultrasound. Cheap models use transistors in avalanche breakdown mode, which is far from being very reliable.

The next stage in any power supply is electrolytes. Fracture of the housing and leakage of electrolyte are not nearly as common as they write on the RuNet, but loss of capacity occurs much more often than failure of active elements. Electrolytic capacitors are checked with a multimeter capable of measuring capacitance. Below the nominal value by 20% or more - we lower the “dead” into the sludge and install a new, good one.

Then there are the active elements. You probably know how to dial diodes and transistors. But there are 2 tricks here. The first is that if a Schottky diode or zener diode is called by a tester with a 12V battery, then the device may show a breakdown, although the diode is quite good. It is better to call these components using a pointer device with a 1.5-3 V battery.

The second is powerful field workers. Above (did you notice?) it is said that their I-Z are protected by diodes. Therefore, powerful field-effect transistors seem to sound like serviceable bipolar transistors, even if they are unusable if the channel is “burnt out” (degraded) not completely.

Here, the only way available at home is to replace them with known good ones, both at once. If there is a burnt one left in the circuit, it will immediately pull a new working one with it. Electronics engineers joke that powerful field workers cannot live without each other. Another prof. joke – “replacement gay couple.” This means that the transistors of the IIN arms must be strictly of the same type.

Finally, film and ceramic capacitors. They are characterized by internal breaks (found by the same tester that checks the “air conditioners”) and leakage or breakdown under voltage. To “catch” them, you need to assemble a simple circuit according to Fig. 7. Step-by-step testing of electrical capacitors for breakdown and leakage is carried out as follows:

• We set on the tester, without connecting it anywhere, the smallest limit for measuring direct voltage (most often 0.2V or 200mV), detect and record the device’s own error;
• We turn on the measurement limit of 20V;
• We connect the suspicious capacitor to points 3-4, the tester to 5-6, and to 1-2 we apply a constant voltage of 24-48 V;
• Switch the multimeter voltage limits down to the lowest;
• If on any tester it shows anything other than 0000.00 (at the very least - something other than its own error), the capacitor being tested is not suitable.

This is where the methodological part of the diagnosis ends and the creative part begins, where all the instructions are based on your own knowledge, experience and considerations.

A couple of impulses

UPSs are a special article due to their complexity and circuit diversity. Here, to begin with, we will look at a couple of samples using pulse width modulation (PWM), which allows us to obtain the best quality UPS. There are a lot of PWM circuits in RuNet, but PWM is not as scary as it is made out to be...

For lighting design

You can simply light the LED strip from any power supply described above, except for the one in Fig. 1, setting the required voltage. SNN with pos. 1 Fig. 3, it’s easy to make 3 of these, for channels R, G and B. But the durability and stability of the LEDs’ glow does not depend on the voltage applied to them, but on the current flowing through them. Therefore, a good power supply for LED strip should include a load current stabilizer; in technical terms - a stable current source (IST).

One of the schemes for stabilizing the light strip current, which can be repeated by amateurs, is shown in Fig. 8. It is assembled on an integrated timer 555 (domestic analogue - K1006VI1). Provides a stable tape current from a power supply voltage of 9-15 V. The amount of stable current is determined by the formula I = 1/(2R6); in this case - 0.7A. The powerful transistor VT3 is necessarily a field-effect transistor; from a draft, due to the base charge, a bipolar PWM simply will not form. Inductor L1 is wound on a ferrite ring 2000NM K20x4x6 with a 5xPE 0.2 mm harness. Number of turns – 50. Diodes VD1, VD2 – any silicon RF (KD104, KD106); VT1 and VT2 – KT3107 or analogues. With KT361, etc. The input voltage and brightness control ranges will decrease.

The circuit works like this: first, the time-setting capacitance C1 is charged through the R1VD1 circuit and discharged through VD2R3VT2, open, i.e. in saturation mode, through R1R5. The timer generates a sequence of pulses with the maximum frequency; more precisely - with a minimum duty cycle. The VT3 inertia-free switch generates powerful impulses, and its VD3C4C3L1 harness smooths them out to direct current.

Note: The duty cycle of a series of pulses is the ratio of their repetition period to the pulse duration. If, for example, the pulse duration is 10 μs, and the interval between them is 100 μs, then the duty cycle will be 11.

The current in the load increases, and the voltage drop across R6 opens VT1, i.e. transfers it from the cut-off (locking) mode to the active (reinforcing) mode. This creates a leakage circuit for the base of VT2 R2VT1+Upit and VT2 also goes into active mode. The discharge current C1 decreases, the discharge time increases, the duty cycle of the series increases and the average current value drops to the norm specified by R6. This is the essence of PWM. At minimum current, i.e. at maximum duty cycle, C1 is discharged through the VD2-R4-internal timer switch circuit.

In the original design, the ability to quickly adjust the current and, accordingly, the brightness of the glow is not provided; There are no 0.68 ohm potentiometers. The easiest way to adjust the brightness is by connecting, after adjustment, a 3.3-10 kOhm potentiometer R* into the gap between R3 and the VT2 emitter, highlighted in brown. By moving its engine down the circuit, we will increase the discharge time of C4, the duty cycle and reduce the current. Another method is to bypass the base junction of VT2 by turning on a potentiometer of approximately 1 MOhm at points a and b (highlighted in red), less preferable, because the adjustment will be deeper, but rougher and sharper.

Unfortunately, to set up this useful not only for IST light tapes, you need an oscilloscope:

1. The minimum +Upit is supplied to the circuit.
2. By selecting R1 (impulse) and R3 (pause) we achieve a duty cycle of 2, i.e. The pulse duration must be equal to the pause duration. You cannot give a duty cycle less than 2!
3. Serve maximum +Upit.
4. By selecting R4, the rated value of a stable current is achieved.

For charging

In Fig. 9 – diagram of the simplest ISN with PWM, suitable for charging a phone, smartphone, tablet (a laptop, unfortunately, will not work) from a homemade solar battery, wind generator, motorcycle or car battery, magneto flashlight “bug” and other low-power unstable random sources power supply See the diagram for the input voltage range, there is no error there. This ISN is indeed capable of producing an output voltage greater than the input. As in the previous one, here there is the effect of changing the polarity of the output relative to the input; this is generally a proprietary feature of PWM circuits. Let's hope that after reading the previous one carefully, you will understand the work of this tiny little thing yourself.

Incidentally, about charging and charging

Charging batteries is a very complex and delicate physical and chemical process, the violation of which reduces their service life several times or tens of times, i.e. number of charge-discharge cycles. The charger must, based on very small changes in battery voltage, calculate how much energy has been received and regulate the charging current accordingly according to a certain law. Therefore, the charger is by no means a power supply, and only batteries in devices with a built-in charge controller can be charged from ordinary power supplies: phones, smartphones, tablets, and certain models of digital cameras. And charging, which is a charger, is a subject for a separate discussion.

Question-remont.ru said:

There will be some sparking from the rectifier, but it's probably not a big deal. The point is the so-called. differential output impedance of the power supply. For alkaline batteries it is about mOhm (milliohms), for acid batteries it is even less. A trance with a bridge without smoothing has tenths and hundredths of an ohm, i.e. approx. 100 – 10 times more. And the starting current of a brushed DC motor can be 6-7 or even 20 times greater than the operating current. Yours is most likely closer to the latter - fast-accelerating motors are more compact and more economical, and the huge overload capacity of the batteries allows you to give the engine as much current as it can handle. for acceleration. A trans with a rectifier will not provide as much instantaneous current, and the engine accelerates more slowly than it was designed for, and with a large slip of the armature. From this, from the large slip, a spark arises, and then remains in operation due to self-induction in the windings.

What can I recommend here? First: take a closer look - how does it spark? You need to watch it in operation, under load, i.e. during sawing.

If sparks dance in certain places under the brushes, it’s okay. My powerful Konakovo drill sparkles so much from birth, and for goodness sake. In 24 years, I changed the brushes once, washed them with alcohol and polished the commutator - that’s all. If you connected an 18V instrument to a 24V output, then a little sparking is normal. Unwind the winding or extinguish the excess voltage with something like a welding rheostat (a resistor of approximately 0.2 Ohm for a power dissipation of 200 W or more), so that the motor operates at the rated voltage and, most likely, the spark will go away. If you connected it to 12 V, hoping that after rectification it would be 18, then in vain - the rectified voltage drops significantly under load. And the commutator electric motor, by the way, doesn’t care whether it is powered by direct current or alternating current.

Specifically: take 3-5 m of steel wire with a diameter of 2.5-3 mm. Roll into a spiral with a diameter of 100-200 mm so that the turns do not touch each other. Place on a fireproof dielectric pad. Clean the ends of the wire until shiny and fold them into “ears”. It is best to immediately lubricate with graphite lubricant to prevent oxidation. This rheostat is connected to the break in one of the wires leading to the instrument. It goes without saying that the contacts should be screws, tightened tightly, with washers. Connect the entire circuit to the 24V output without rectification. The spark is gone, but the power on the shaft has also dropped - the rheostat needs to be reduced, one of the contacts needs to be switched 1-2 turns closer to the other. It still sparks, but less - the rheostat is too small, you need to add more turns. It is better to immediately make the rheostat obviously large so as not to screw on additional sections. It’s worse if the fire is along the entire line of contact between the brushes and the commutator or spark tails trail behind them. Then the rectifier needs an anti-aliasing filter somewhere, according to your data, from 100,000 µF. Not a cheap pleasure. The “filter” in this case will be an energy storage device for accelerating the motor. But it may not help if the overall power of the transformer is not enough. Efficiency of brushed DC motors is approx. 0.55-0.65, i.e. trans is needed from 800-900 W. That is, if the filter is installed, but still sparks with fire under the entire brush (under both, of course), then the transformer is not up to the task. Yes, if you install a filter, then the diodes of the bridge must be rated for triple the operating current, otherwise they may fly out from the surge of charging current when connected to the network. And then the tool can be launched 5-10 seconds after being connected to the network, so that the “banks” have time to “pump up”.

And the worst thing is if the tails of sparks from the brushes reach or almost reach the opposite brush. This is called all-round fire. It very quickly burns out the collector to the point of complete disrepair. There can be several reasons for a circular fire. In your case, the most probable is that the motor was turned on at 12 V with rectification. Then, at a current of 30 A, the electrical power in the circuit is 360 W. The anchor slides more than 30 degrees per revolution, and this is necessarily a continuous all-round fire. It is also possible that the motor armature is wound with a simple (not double) wave. Such electric motors are better at overcoming instantaneous overloads, but they have a starting current - mother, don’t worry. I can’t say more precisely in absentia, and there’s no point in it – there’s hardly anything we can fix here with our own hands. Then it will probably be cheaper and easier to find and purchase new batteries. But first, try turning on the engine at a slightly higher voltage through the rheostat (see above). Almost always, in this way it is possible to shoot down a continuous all-round fire at the cost of a small (up to 10-15%) reduction in power on the shaft.

Circuit diagram of a very simple, high-power, high-efficiency switching adjustable voltage regulator

Good afternoon, dear radio amateurs!
Welcome to the website ““

Today we are with you Let's consider the circuit of a powerful pulse adjustable voltage stabilizer. This circuit can be used both for installation in amateur radio devices with a fixed output voltage, and in power supplies with an adjustable output voltage. Although the circuit is very simple, it has fairly good characteristics and can be repeated by radio amateurs with any basic training.

The basis of this stabilizer is a specialized microcircuit LM-2596T-ADJ, which is precisely intended for the construction of pulse regulators of adjustable voltage. The microcircuit has built-in output current protection and thermal protection. In addition, the circuit contains a diode D1 – Schottky diode type 1N5822 And throttle factory-made (in principle, you can make it yourself) inductance 120 microhenry. Capacitors C1 and C2 - for an operating voltage of at least 50 volts, resistor R1 with a power of 0.25 watts.

To obtain an adjustable output voltage, it is necessary to connect a variable resistor to pins 1 and 2 (with the shortest possible length of connection wires). If it is necessary to obtain a fixed voltage at the output, then instead of a variable resistor, a constant one is installed, the value of which is selected experimentally.

In addition, the LM-2596 series has fixed stabilizers for voltages of 3.3 V, 5 V and 12 V, the connection diagram of which is even simpler (can be viewed in the datasheet).

Specifications:

As you can see, the characteristics for using this circuit in a power supply are quite decent (according to the datasheet, the output voltage is regulated within 1.2-37 volts). The efficiency of the stabilizer at an input voltage of 12 volts, an output voltage of 3 volts and a load current of 3 amperes is 73%. When making this stabilizer, we must not forget that the higher the input voltage and the lower the output voltage, the permissible load current will decrease, so this stabilizer must be installed on a radiator with an area of ​​at least 100 sq.cm. If the circuit will operate at low load currents, then it is not necessary to install a radiator.

Below are the appearance of the main parts, their approximate cost in online stores and the location of the parts on the board.

Based on the layout of the parts, making a printed circuit board yourself is not difficult.

This circuit can operate in output current stabilization mode, which allows it to be used to charge batteries, power a powerful LED or a group of powerful LEDs, etc.

To turn the circuit into current stabilization mode, it is necessary to install a resistor in parallel with resistor R1, the value of which is determined by the formula: R = 1.23/I

The cost of this scheme is approximately 300 rubles, which is at least 100 rubles cheaper than buying a finished product.

The operation of almost any electronic circuit requires the presence of one or more constant voltage sources, and in the vast majority of cases a stabilized voltage is used. Stabilized power supplies use either linear or switching stabilizers. Each type of converter has its own advantages and, accordingly, its own niche in power supply circuits. The undoubted advantages of switching stabilizers include higher efficiency values, the ability to obtain high output current values ​​and high efficiency with a large difference between the input and output voltages.

The operating principle of a buck pulse stabilizer

Figure 1 shows a simplified diagram of the power section of the IPSN.

Rice. 1.

Field effect transistor VT performs high-frequency current switching. In pulse stabilizers, the transistor operates in switching mode, that is, it can be in one of two stable states: full conduction and cutoff. Accordingly, the operation of the IPSN consists of two alternating phases - the energy pumping phase (when the VT transistor is open) and the discharge phase (when the transistor is closed). The operation of the IPSN is illustrated in Figure 2.

Rice. 2. Operating principle of IPSN: a) pumping phase; b) discharge phase; c) timing diagrams

The energy pumping phase continues throughout the time interval T I. During this time, the switch is closed and conducts current I VT. Next, the current passes through the inductor L to the load R, shunted by the output capacitor C OUT. In the first part of the phase, the capacitor supplies current I C to the load, and in the second half, it takes part of the current I L from the load. The magnitude of the current I L continuously increases, and energy is accumulated in the inductor L, and in the second part of the phase - on the capacitor C OUT. The voltage across the diode V D is equal to U IN (minus the voltage drop across the open transistor), and the diode is closed during this phase - no current flows through it. The current I R flowing through the load R is constant (the difference I L - I C), accordingly, the voltage U OUT at the output is also constant.

The discharge phase occurs during the time T P: the switch is open and no current flows through it. It is known that the current flowing through the inductor cannot change instantly. The current IL, constantly decreasing, flows through the load and closes through the diode V D. In the first part of this phase, the capacitor C OUT continues to accumulate energy, taking part of the current I L from the load. In the second half of the discharge phase, the capacitor also begins to supply current to the load. During this phase, the current I R flowing through the load is also constant. Therefore, the output voltage is also stable.

Main settings

First of all, we note that according to their functional design, they distinguish between IPSN with adjustable and fixed output voltage. Typical switching circuits for both types of IPSN are presented in Figure 3. The difference between them is that in the first case, the resistor divider, which determines the value of the output voltage, is located outside the integrated circuit, and in the second, inside. Accordingly, in the first case, the value of the output voltage is set by the user, and in the second, it is set during the manufacture of the microcircuit.

Rice. 3. Typical switching circuit for IPSN: a) with adjustable and b) with fixed output voltage

The most important parameters of IPSN include:

• Range of permissible input voltage values ​​U IN_MIN…U IN_MAX.
• The maximum value of the output current (load current) I OUT_MAX.
• Nominal value of the output voltage U OUT (for IPSN with a fixed output voltage value) or range of output voltage values ​​U OUT_MIN ...U OUT_MAX (for IPSN with an adjustable output voltage value). Often reference materials indicate that the maximum value of the output voltage U OUT_MAX is equal to the maximum value of the input voltage U IN_MAX. In reality this is not entirely true. In any case, the output voltage is less than the input voltage, at least by the amount of voltage drop across the key transistor U DROP. With an output current value equal to, for example, 3A, the value of U DROP will be 0.1...1.0V (depending on the selected IPSN microcircuit). Approximate equality of U OUT_MAX and U IN_MAX is possible only at very low load current values. Note also that the process of stabilizing the output voltage itself involves a loss of several percent of the input voltage. The declared equality of U OUT_MAX and U IN_MAX should be understood only in the sense that there are no other reasons for reducing U OUT_MAX other than those indicated above in a specific product (in particular, there are no explicit restrictions on the maximum value of the fill factor D). The value of the feedback voltage U FB is usually indicated as U OUT_MIN. In reality, U OUT_MIN should always be several percent higher (for the same stabilization reasons).
• Accuracy of output voltage setting. Set as a percentage. It makes sense only in the case of IPSN with a fixed output voltage value, since in this case the voltage divider resistors are located inside the microcircuit, and their accuracy is a parameter controlled during manufacturing. In the case of IPSN with an adjustable output voltage value, the parameter loses its meaning, since the accuracy of the divider resistors is selected by the user. In this case, we can only talk about the magnitude of the output voltage fluctuations relative to a certain average value (the accuracy of the feedback signal). Let us recall that in any case, this parameter for switching voltage stabilizers is 3...5 times worse compared to linear stabilizers.
• Voltage drop across open transistor R DS_ON. As already noted, this parameter is associated with an inevitable decrease in the output voltage relative to the input voltage. But something else is more important - the higher the resistance value of the open channel, the more energy is dissipated in the form of heat. For modern IPSN microcircuits, values ​​up to 300 mOhm are a good value. Higher values ​​are typical for chips developed at least five years ago. Note also that the value of R DS_ON is not a constant, but depends on the value of the output current I OUT.
• Duty cycle duration T and switching frequency F SW. The duration of the working cycle T is determined as the sum of the intervals T I (pulse duration) and T P (pause duration). Accordingly, the frequency F SW is the reciprocal of the operating cycle duration. For some part of the IPSN, the switching frequency is a constant value determined by the internal elements of the integrated circuit. For another part of the IPSN, the switching frequency is set by external elements (usually an external RC circuit), in this case the range of permissible frequencies F SW_MIN ... F SW_MAX is determined. A higher switching frequency allows the use of chokes with a lower inductance value, which has a positive effect on both the dimensions of the product and its price. Most ISPS use PWM control, that is, the T value is constant, and during the stabilization process the T I value is adjusted. Pulse frequency modulation (PFM control) is used much less frequently. In this case, the value of T I is constant, and stabilization is carried out by changing the duration of the pause T P. Thus, the values ​​of T and, accordingly, F SW become variable. In reference materials in this case, as a rule, a frequency is specified corresponding to a duty cycle equal to 2. Note that the frequency range F SW_MIN ...F SW_MAX of an adjustable frequency should be distinguished from the tolerance gate for a fixed frequency, since the tolerance value is often indicated in reference materials manufacturer.
• Duty factor D, which is equal to the percentage
  the ratio of T I to T. Reference materials often indicate “up to 100%”. Obviously, this is an exaggeration, since if the key transistor is constantly open, then there is no stabilization process. In most models released on the market before approximately 2005, due to a number of technological limitations, the value of this coefficient was limited above 90%. In modern IPSN models, most of these limitations have been overcome, but the phrase “up to 100%” should not be taken literally.
• Efficiency factor (or efficiency). As is known, for linear stabilizers (fundamentally step-down) this is the percentage ratio of the output voltage to the input, since the values ​​of the input and output current are almost equal. For switching stabilizers, the input and output currents can differ significantly, so the percentage ratio of output power to input power is taken as efficiency. Strictly speaking, for the same IPSN microcircuit, the value of this coefficient can differ significantly depending on the ratio of the input and output voltages, the amount of current in the load and the switching frequency. For most IPSN, maximum efficiency is achieved at a load current value of the order of 20...30% of the maximum permissible value, so the numerical value is not very informative. It is more advisable to use the dependence graphs that are provided in the manufacturer’s reference materials. Figure 4 shows efficiency graphs for a stabilizer as an example. . Obviously, using a high-voltage stabilizer at low actual input voltage values ​​is not a good solution, since the efficiency value drops significantly as the load current approaches its maximum value. The second group of graphs illustrates the more preferable mode, since the efficiency value weakly depends on fluctuations in the output current. The criterion for the correct choice of a converter is not so much the numerical value of the efficiency, but rather the smoothness of the graph of the function of the current in the load (the absence of a “blockage” in the region of high currents).

Rice. 4.

The given list does not exhaust the entire list of IPSN parameters. Less significant parameters can be found in the literature.

Special Features
pulse voltage stabilizers

In most cases, IPSN have a number of additional functions that expand the possibilities of their practical application. The most common are the following:

• The “On/Off” or “Shutdown” load shutdown input allows you to open the key transistor and thus disconnect the voltage from the load. As a rule, it is used for remote control of a group of stabilizers, implementing a certain algorithm for applying and turning off individual voltages in the power supply system. In addition, it can be used as an input for emergency power off in case of an emergency.
• Normal state output “Power Good” is a generalizing output signal confirming that the IPSN is in normal operating condition. The active signal level is formed after the completion of transient processes from the supply of input voltage and, as a rule, is used either as a sign of the serviceability of the ISPN, or to trigger the following ISPN in serial power supply systems. The reasons why this signal can be reset: the input voltage drops below a certain level, the output voltage goes beyond a certain range, the load is turned off by the Shutdown signal, the maximum current value in the load is exceeded (in particular, the fact of a short circuit), temperature shutdown of the load and some other. The factors that are taken into account when generating this signal depend on the specific IPSN model.
• The external synchronization pin “Sync” provides the ability to synchronize the internal oscillator with an external clock signal. Used to organize joint synchronization of several stabilizers in complex power supply systems. Note that the frequency of the external clock signal does not have to coincide with the natural frequency of the FSW, however, it must be within the permissible limits specified in the manufacturer’s materials.
• The Soft Start function provides a relatively slow increase in output voltage when voltage is applied to the input of the IPSN or when the Shutdown signal is turned on at the falling edge. This function allows you to reduce current surges in the load when the microcircuit is turned on. The operating parameters of the soft start circuit are most often fixed and determined by the internal components of the stabilizer. Some IPSN models have a special Soft Start output. In this case, the startup parameters are determined by the ratings of external elements (resistor, capacitor, RC circuit) connected to this pin.
• Temperature protection is designed to prevent chip failure if the crystal overheats. An increase in the temperature of the crystal (regardless of the reason) above a certain level triggers a protective mechanism - a decrease in the current in the load or its complete shutdown. This prevents further rise in die temperature and damage to the chip. Returning the circuit to voltage stabilization mode is possible only after the microcircuit has cooled. Note that temperature protection is implemented in the vast majority of modern IPSN microcircuits, but a separate indication of this particular condition is not provided. The engineer will have to guess for himself that the reason for the load shutdown is precisely the operation of the temperature protection.
• Current protection consists of either limiting the amount of current flowing through the load or disconnecting the load. The protection is triggered if the load resistance is too low (for example, there is a short circuit) and the current exceeds a certain threshold value, which can lead to failure of the microcircuit. As in the previous case, diagnosing this condition is the concern of the engineer.

One last note regarding the parameters and functions of the IPSN. In Figures 1 and 2 there is a discharge diode V D. In fairly old stabilizers, this diode is implemented precisely as an external silicon diode. The disadvantage of this circuit solution was the high voltage drop (approximately 0.6 V) across the diode in the open state. Later designs used a Schottky diode, which had a voltage drop of approximately 0.3 V. In the last five years, designs have used these solutions only for high-voltage converters. In most modern products, the discharge diode is made in the form of an internal field-effect transistor operating in antiphase with the key transistor. In this case, the voltage drop is determined by the resistance of the open channel and at low load currents gives an additional gain. Stabilizers using this circuit design are called synchronous. Please note that the ability to operate from an external clock signal and the term “synchronous” are not related in any way.

with low input voltage

Considering the fact that in the STMicroelectronics range there are approximately 70 types of IPSN with a built-in key transistor, it makes sense to systematize all the diversity. If we take as a criterion a parameter such as the maximum value of the input voltage, then four groups can be distinguished:

1. IPSN with low input voltage (6 V or less);

2. IPSN with input voltage 10...28 V;

3. IPSN with input voltage 36…38 V;

4. IPSN with high input voltage (46 V and above).

The parameters of stabilizers of the first group are given in Table 1.

Table 1. IPSN with low input voltage

Name	Exit current, A	Input voltage, V		Day off voltage, V		Efficiency, %	Switching frequency, kHz		Functions and flags
	I OUT	V IN		V OUT		h	FSW	R DSON	On/Off	Sync. Pin	Soft Start	Pow Good
	Max	Min	Max	Min	Max	Max	Type
L6925D	0,8	2,7	5,5	0,6	5,5	95	600	240	+	+	+	+
L6926	0,8	2,0	5,5	0,6	5,5	95	600	240	+	+	+	+
L6928	0,8	2,0	5,5	0,6	5,5	95	1450	240	+	+	+	+
PM8903A	3,0	2,8	6,0	0,6	6,0	96	1100	35	+	+	+	+
ST1S06A	1,5	2,7	6,0	0,8	5,0	92	1500	150	+	–	+	–
ST1S09	2,0	4,5	5,5	0,8	5,0	95	1500	100	*	–	+	+
ST1S12	0,7	2,5	5,5	0,6	5,0	92	1700	250	+		+	–
ST1S15	0,5	2,3	5,5	Fix. 1.82 and 2.8 V		90	6000	350	+	–	+	–
ST1S30	3,0	2,7	6,0	0,8	5,0	85	1500	100	*	–	+	+
ST1S31	3,0	2,8	5,5	0,8	5,5	95	1500	60	+	–	+	–
ST1S32	4,0	2,8	5,5	0,8	5,5	95	1500	60	+	–	+	–
* – the function is not available for all versions.

Back in 2005, the line of stabilizers of this type was incomplete. It was limited to microcircuits. These microcircuits had good characteristics: high accuracy and efficiency, no restrictions on the duty cycle value, the ability to adjust the frequency when operating from an external clock signal, and an acceptable RDSON value. All this makes these products in demand today. A significant drawback is the low maximum output current. There were no stabilizers for load currents of 1 A and higher in the line of low-voltage IPSN from STMicroelectronics. Subsequently, this gap was eliminated: first, stabilizers for 1.5 and 2 A ( and ) appeared, and in recent years - for 3 and 4 A ( , And ). In addition to increasing the output current, the switching frequency has increased and the open channel resistance has decreased, which has a positive effect on the consumer properties of the final products. We also note the emergence of IPSN microcircuits with a fixed output voltage ( and ) - there are not very many such products in the STMicroelectronics line. The latest addition, with an RDSON value of 35 mOhm, is one of the best in the industry, which, combined with extensive functionality, promises good prospects for this product.

The main application area for products of this type is battery-powered mobile devices. A wide input voltage range ensures stable operation of the equipment at different battery charge levels, and high efficiency minimizes the conversion of input energy into heat. The latter circumstance determines the advantages of switching stabilizers over linear ones in this area of ​​user applications.

In general, this group of STMicroelectronics is developing quite dynamically - approximately half of the entire line has appeared on the market in the last 3-4 years.

Switching buck stabilizers
with input voltage 10…28 V

The parameters of the converters of this group are given in Table 2.

Table 2. IPSN with input voltage 10…28 V

Name	Exit current, A	Input voltage, V		Day off voltage, V		Efficiency, %	Switching frequency, kHz	Open channel resistance, mOhm	Functions and flags
	I OUT	V IN		V OUT		h	FSW	R DSON	On/Off	Sync. Pin	Soft Start	Pow Good
	Max	Min	Max	Min	Max	Max	Type
L5980	0,7	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5981	1,0	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5983	1,5	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5985	2,0	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5986	2,5	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5987	3,0	2,9	18,0	0,6	18,0	93	250…1000	140	+	+	+	–
L5988D	4,0	2,9	18,0	0,6	18,0	95	400…1000	120	+	+	+	–
L5989D	4,0	2,9	18,0	0,6	18,0	95	400…1000	120	+	–	+	+
L7980	2,0	4,5	28,0	0,6	28,0	93	250…1000	160	+	+	+	–
L7981	3,0	4,5	28,0	0,6	28,0	93	250…1000	160	+	+	+	–
ST1CC40	2,0	3,0	18,0	0,1	18,0	n.d.	850	95	+	–	+	–
ST1S03	1,5	2,7	16,0	0,8	12,0	79	1500	280	–	–	+	–
ST1S10	3,0	2,7	18,0	0,8	16,0	95	900	120	+	+	+	–
ST1S40	3,0	4,0	18,0	0,8	18,0	95	850	95	+	–	+	–
ST1S41	4,0	4,0	18,0	0,8	18,0	95	850	95	+	–	+	–
ST763AC	0,5	3,3	11,0	Fix. 3.3		90	200	1000	+	–	+	–

Eight years ago this group was represented only by microcircuits , and with input voltage up to 11 V. The range from 16 to 28 V remained empty. Of all the listed modifications, only , but the parameters of this IPSN poorly correspond to modern requirements. We can assume that during this time the nomenclature of the group under consideration has been completely updated.

Currently, the base of this group is microcircuits . This line is designed for the entire range of load currents from 0.7 to 4 A, provides a full set of special functions, the switching frequency is adjustable within a fairly wide range, there are no restrictions on the duty cycle, the efficiency and open-channel resistance values ​​meet modern requirements. There are two significant disadvantages in this series. Firstly, there is no built-in discharge diode (except for microcircuits with the D suffix). The accuracy of output voltage regulation is quite high (2%), but the presence of three or more external elements in the feedback compensation circuit cannot be considered an advantage. The microcircuits differ from the L598x series only in a different input voltage range, but the circuit design, and, consequently, the advantages and disadvantages are similar to the L598x family. As an example, Figure 5 shows a typical connection circuit for a three-amp microcircuit. There is also a discharge diode D and compensation circuit elements R4, C4 and C5. The F SW and SYNCH inputs remain free, therefore, the converter operates from an internal oscillator with the default frequency F SW.

This review is dedicated to the switching stabilizer module, which is offered by online stores under the name "5A Lithium Charger CV CC Buck Step Down Power Module LED Driver". Thus, the module is a pulse-step-down converter designed for charging lithium-ion batteries in CV (constant voltage) and CC (constant current) modes, as well as for powering LEDs. This device costs about 2 USD. Structurally, the module is a printed circuit board on which all elements are installed, including signal LEDs and adjustment controls. The appearance of the module is shown in Fig. 1.

A drawing of the printed circuit board is shown in Fig. 2.

According to the manufacturer's specifications, the module has the following technical characteristics:

• Input voltage 6-38V DC.
• Output voltage adjustable 1.25-36 VDC.
• Output current 0-5 A (adjustable).
• Load power up to 75 VA.
• Efficiency is more than 96%.
• There is built-in protection against overheating and short circuit in the load.
• Module dimensions 61.7x26.2x15 mm.
• Weight 20 grams.

The combination of low price, small size and high technical characteristics aroused the author's interest and desire to experimentally determine the main characteristics of the module.
The manufacturer does not provide an electrical circuit diagram, so I had to draw it myself. The result of this work is presented in Fig. 3.

The basis of the device is the DA2 XL4015 chip, which is an original Chinese design. This chip is very similar to the popular LM2596, but has improved characteristics. Apparently this is achieved by using a powerful field-effect transistor as a power switch. The description of this microcircuit is given in L1. In this device, the microcircuit is included in full accordance with the manufacturer's recommendations. The variable resistor “CV” is the output voltage regulator. The adjustable output current limiting circuit is based on the DA3.1 operational amplifier. This amplifier compares the voltage drop across current sense resistor R9 with the regulated voltage across variable resistor “CC”. Using this resistor, you can set the desired level of current limitation in the stabilizer load.

If the specified current value is exceeded, a high-level signal will appear at the output of the amplifier, the red HL2 LED will open and the voltage at input 2 of the DA2 chip will increase, which will lead to a decrease in the voltage and current at the output of the stabilizer. In addition, the glow of HL2 will indicate that the module is operating in current stabilization (CC) mode. Capacitor C5 must ensure the stability of the current control unit.

The second operational amplifier DA3.2 contains a signaling device for reducing the current in the load to a value less than 9% of the specified maximum current. If the current exceeds the specified value, then the blue LED HL3 lights up, otherwise the green LED HL1 lights up. When charging lithium-ion batteries, a decrease in the charging current is one of the signs that charging has ended.
The DA1 chip contains a stabilizer with an output voltage of 5V. This voltage is used to power the DA3 operational amplifier, and it is also used to form the reference voltage for the current limiter and the current low alarm.

The voltage drop across the current-measuring resistor is not compensated in any way; therefore, as the current in the load increases, the output voltage of the stabilizer decreases. To reduce this drawback, the value of the current measuring resistor is chosen to be quite small (0.05 Ohm). Because of this, drift in the DA3 operational amplifier can cause noticeable instability in both the output current limiting level and the alarm level.
Tests of the module have shown that the output resistance of the stabilizer in voltage regulation (CV) mode is almost completely determined by the current measuring resistor and is about 0.06 Ohm.
Voltage stabilization factor is about 400.
To evaluate the heat dissipation, a voltage of 12V was applied to the module input. The output voltage was set to 5V with a load resistance of 2.5 Ohms (current 2A). After 30 minutes, the DA2 chip, inductor L1 and diode VD1 heated up to 71, 64 and 48 degrees Celsius, respectively.

Operation in load current stabilization mode (SS) was accompanied by the DA2 microcircuit transitioning to the pulse burst generation mode. The repetition frequency and duration of the bursts varied within wide limits depending on the magnitude of the current. In this case, the effect of current stabilization took place, but the ripples at the module output increased significantly. In addition, operation of the device in CC mode was accompanied by a rather loud squeak, the source of which was inductor L1.
The operation of the current reduction alarm did not raise any complaints. The module successfully withstood a short circuit in the load.

Thus, the module is operational in both CV and CC modes, but when using it, the above-described features should be taken into account.
This review is written based on the results of a study of one copy of the device, which makes the results obtained purely indicative.
According to the author, the described switching stabilizer can be successfully used if a cheap, compact power source with satisfactory characteristics is required.

List of radioelements

Designation	Type	Denomination	Quantity	Note	Shop	My notepad
DA1	Linear regulator	LM317L	1			To notepad
DA2	Chip	XL4015	1			To notepad
DA3	Operational amplifier	LM358	1			To notepad
VD1	Schottky diode	SK54	1			To notepad
HL1	Light-emitting diode	Green	1			To notepad
HL2	Light-emitting diode	Red	1			To notepad
HL3	Light-emitting diode	Blue	1			To notepad
C1, C6	Electrolytic capacitor	220 µF 50 V	2			To notepad
C2-C4, C7	Capacitor	0.47 µF	4			To notepad
C5	Capacitor	0.01 µF	1			To notepad
R1	Resistor	680 Ohm	1			To notepad
R2	Resistor	220 Ohm	1			To notepad
R3	Resistor	330 Ohm	1			To notepad
R4	Resistor	18 kOhm	1			To notepad
R7	Resistor	100 kOhm	1			To notepad
R8	Resistor	10 kOhm	1